This invention relates to an RF band reject filter, to a base station power amplifier for a cellular radio network, to a duplexer for a mobile telephone handset, and to a low noise amplifier (LNA) input stage.
Surface acoustic wave (SAW) devices have been studied and gradually commercialised since the mid 1960s. Such devices typically have electrodes in the form of interlocked xe2x80x9cfingersxe2x80x9d (so-called inter-digital electrodes) formed on a piezoelectric substrate. When high frequency signals are applied to the input electrodes, mechanical vibrations in the form of travelling acoustic waves are generated in the piezoelectric substrate which are picked up by the output electrodes. Generally speaking, when the wavelength of the surface acoustic waves and the period of the electrode xe2x80x9cfingersxe2x80x9d are the same, the magnitude of the surface acoustic waves are at their greatest and the device has a low electrical impedance. At other input frequencies, the device appears to have a higher electrical impedance.
Thus, such a so-called xe2x80x9cSAW resonator filterxe2x80x9d can be made to have a very precise and narrow (typically having a Q factor over 1000), band pass characteristic. Furthermore, since surface acoustic waves travel across the substrate 100000 times more slowly than the speed of electromagnetic waves, such devices are generally compact. In practice, such devices can be used in a ladder configuration (with a plurality of shunt and plurality of series resonator filters used together). This allows a combined band pass characteristic to be tuned as desired.
Thus such devices have found many uses. However, such devices suffer from two significant disadvantages which prevents their use in some applications. Firstly, band pass filters produced using SAW resonators typically have relatively high insertion losses typically of a minimum of 1 or 2 dB. The state of the art presently is an insertion loss of about 1 dB in the pass band with a rejection of about 15 dB in the stop band for a single stage band pass ladder filter. The losses typically occur as a result of visco-elastic attenuations and/or mode conversions from SAW to bulk acoustic waves when the electrical energy is converted to acoustic energy and travels around the SAW filter cavity. Secondly, the power handling capability of SAW filters is limited. At high powers, the ultrasonic vibration to which the metallic electrodes are subjected eventually causes the metal grain boundaries to migrate. Thus, for example, at the present 1800, 1900 and 2100 MHz cellular mobile bands, such filters cannot be used for a mobile handset duplexer because at these frequencies, such filters cannot survive for a realistic length of time at the desired power levels of approximately 30 dBm.
Relatively little work has been done on SAW notch or band reject filters to date. Of the little work which has been reported, most of it has focussed on the development of narrowband notch filters. One of the first publications on SAW notch filters was in U.S. Pat. No. 4,577,168 (Hartman). Various techniques for implementing SAW notch filters are described in which the conductance within the passband of a Single Phase Unidirectional Transducer (SPUDT) SAW transducer was used as an impedance element to create a notch filter. One implementation used the impedance of a SPUDT in conjunction with an RF transformer and other implementations consisted of replacing the capacitors in a bridge-T type notch filter with a SPUDT transducer impedance element. This approach has one disadvantage in that SPUDT transducers fall into the class of Finite Impulse Response Devices and hence the device must be made longer if narrow notch bandwidths are to be achieved. Furthermore, SPUDT type devices are not easily manufactured at elevated frequencies since xe2x85x9 wavelength electrodes are required.
A variation of this technique is described in S. Gopani and B. A. Horine xe2x80x9cSAW Waveguide-Coupled Resonator Notch Filterxe2x80x9d, Ultrasonics Symposium, 1990, in which a Two-Pole Waveguide Coupled (WGC) Resonator is embedded in an all pass network to implement a notch filter. This technique has two major disadvantages. Firstly the WGC resonator is limited to Quartz hence only bandwidth of 0.1% are attainable and secondly, the resonator has a very poor shape factor of around 5.3 since a typical device might have a 40 dB stopband width of 84 kHz and the 3 dB stopband width of 444 kHz. The device described had a centre frequency of 247 MHz and the insertion losses in the passband were of the order of 4 dB.
A further modification is described in P. A. Lorenz and D. F. Thompson, xe2x80x9cWide Bandwidth Low Cost SAW Notch Filtersxe2x80x9d, Ultrasonics Symposium, 1998. This technique consisted of placing two single pole SAW resonators in series with a resonator in between them. This technique achieved notch depths of more than 40 dB but had a relatively poor shape factor of 4.3 where the 40 dB stopband width was 86 kHz and the 3 dB stopband width was 370 kHz at a centre frequency of 420 MHz. Insertion losses in the stop band were approximately 5 dB or less.
Other simpler implementations consist of using a single pole SAW resonator in series with the signal to obtain a notch at the anti-resonance frequency. Although simple, this filter has a relatively narrow rejection bandwidth, and the shape factor is very poor.
All the reported SAW notch filter developments focused on narrow band notch filters versus wider band reject filters. Furthermore, the techniques consisted of using the impedance of a SAW SPUDT or resonator device in an all pass network to generate a notch response near the passband of the SAW device and leveraged the capacitive properties of the SAW device away from the notch to form an all pass network. Rather poor notch shape factors and insertion losses have been achieved in the reported literature. Therefore there is a need for wider rejection band devices and or lower insertion losses within the passband.
According to a first aspect of the invention there is provided an RF band reject filter comprising a shunt acoustic resonator and a series acoustic resonator, the shunt resonator being arranged to resonate generally at the reject frequency band and the series resonator being arranged to be anti-resonant generally at the reject frequency band.
The precise placement of the series resonator anti-resonance frequency and shunt resonator resonance frequency can be adjusted to achieve a desired tradeoff between rejection band depth or rejection band width as explained in detail below.
This arrangement overcomes the power handling problem. Conventionally, acoustic resonators are configured to provide a band pass configuration. In this configuration, the series resonators are chosen to be at resonance in the pass band and the shunt resonators are chosen to be at anti-resonance in the pass band. However, by reversing this configuration, a notch or band reject filter is produced.
Such a band reject filter could be used, for example, at a power amplifier output in a base station or handset transceiver with the reject band being tuned to the receive band noise. The filter will readily allow a large transmit signal to pass since the filter is tuned to the receive band and it is thus at receive frequencies that strong acoustic resonance occurs. Since the receive band power is low, and since in the transmit band there is virtually no acoustic resonance, acoustic-electric migration is not a significant factor in such a filter. Thus, the power handling of such a device in the transmit band is limited by electro-migration (and eventually arcing between the electrodes). This occurs at much higher power levels than the power levels which may be passed by a prior art band pass acoustic resonator filter at its pass band frequency.
Furthermore, in this band reject configuration of the present invention, the filter appears as a high Q series and shunt capacitor outside its reject band. By including a high Q matching network, the capacitance (outside the reject band) can be substantially reduced so that insertion losses in the region of 0.2 dB are attainable for a filter having 15 dB attenuation in the reject band.
Thus, for example, such a filter may be used in an UMTS base station in order to reduce the cost of the duplexer module (which presently makes up about 10% of the total cost of the base station cabinet).
Thus, in a second aspect, there is provided a base station power amplifier for a cellular radio network, the power amplifier including at least one inter-stage band reject filter comprising a shunt acoustic resonator and a series acoustic resonator, the shunt resonator being arranged to resonate generally at the reject frequency band and the series resonator being arranged to anti-resonant generally at the reject frequency band.
The use of the band reject filter of the present invention as an interstage filter for a power amplifier is particularly suitable since as noted above, the band reject filter of the present invention appears as a high Q series and shunt capacitor outside the rejection band. The equivalent capacitor has a short delay and therefore produces a very wideband characteristic with good phase linearity and amplitude flatness. This is in contrast to pass band resonators which have a parabolic group delay and hence poor phase linearity. Furthermore, such pass band resonator filters are (as noted above) too lossy in the pass band and have insufficient amplitude flatness for a cellular radio base station amplifier.
In another aspect, there is provided an LNA input stage including a band reject filter comprising a shunt acoustic resonator and a series acoustic resonator, the shunt resonator being arranged to resonate generally at the reject frequency band and the series resonator being arranged to anti-resonant generally at the reject frequency band. In this configuration, forming part of an LNA input stage, the filter may be used to reject power in the transmit band which is incident on the LNA.
In this way (as explained below), the transmit/receive isolation requirements for the base station duplexer may be reduced, for example, from of the order of 90 dB to of the order of 45 dB. This allows a significantly cheaper duplexer to be used for the base station.
Further, another aspect of the invention provides a duplexer for a mobile telephone handset including an RF band reject filter comprising a shunt acoustic resonator and a series acoustic resonator, the shunt resonator being arranged to resonate generally at the reject frequency band and the series resonator being arranged to anti-resonant generally at the reject frequency band.
Such filters may also be used to reject aircraft band interference, TV or radio signals in a handset or base station. They may also be used to suppress an unwanted LO, sideband or image frequency in a TX or RX chain, where the distortion or loss introduced by a bandpass filter would be unacceptable. Such a situation might arise in the case of a transmitter which must carry a predistorted signal such as would be the case in a system employing a baseband predistortion power amplifier. Since any filters would be within the correction loop, they must provide very little amplitude or phase distortion over the frequency ranges of interest.
Furthermore, since the equivalent circuit for a SAW resonator and a thin film bulk acoustic resonator (for example, the FBAR product available from Agilent) are identical, the invention may use a SAW resonator or a thin bulk acoustic resonator for the acoustic resonator component. One advantage of the thin film bulk acoustic resonator implementation is that the filter may be implemented completely in silicon which may, in some applications, provide packaging advantages over a SAW resonator configuration.
Other aspects and features of the present invention will become apparent to those ordinarily skilled in the art upon review of the following description of specific embodiments of the invention in conjunction with the accompanying figures.